Systems and methods for auto-calibration of resistive temperature sensors

ABSTRACT

The invention relates to systems and methods for calibrating and using resistance temperature detectors. In one embodiment, the system includes a calibration circuit comprising a resistance temperature detector in a bridge circuit with at least one potentiometer, and a programmable gain amplifier coupled to the bridge circuit. Embodiments of the invention further comprise methods for calibrating the bridge circuit and the programmable gain amplifier for use with the resistance temperature detector and methods for determining the self heating voltage of the bridge circuit.

BACKGROUND

1. Field of Invention

The present invention relates to microfluidic devices and temperaturecontrol of the microfluidic devices for performing biological reactions.In some embodiments, the present invention relates to systems andmethods for calibrating and using a resistance temperature detector foruse in a microfluidic device.

2. Discussion of the Background

The detection of nucleic acids is central to medicine, forensic science,industrial processing, crop and animal breeding, and many other fields.The ability to detect disease conditions (e.g., cancer), infectiousorganisms (e.g., HIV), genetic lineage, genetic markers, and the like,is ubiquitous technology for disease diagnosis and prognosis, markerassisted selection, identification of crime scene features, the abilityto propagate industrial organisms and many other techniques.Determination of the integrity of a nucleic acid of interest can berelevant to the pathology of an infection or cancer.

One of the most powerful and basic technologies to detect smallquantities of nucleic acids is to replicate some or all of a nucleicacid sequence many times, and then analyze the amplification products.Polymerase chain reaction (PCR) is a well-known technique for amplifyingDNA. With PCR, one can produce millions of copies of DNA starting from asingle template DNA molecule. PCR includes phases of “denaturation,”“annealing,” and “extension.” These phases are part of a cycle which isrepeated a number of times so that at the end of the process there areenough copies to be detected and analyzed. For general detailsconcerning PCR, see Sambrook and Russell, Molecular Cloning—A LaboratoryManual (3rd Ed.), Vols. 1 -3, Cold Spring Harbor Laboratory, Cold SpringHarbor, N.Y. (2000); Current Protocols in Molecular Biology, F. M.Ausubel et al., eds., Current Protocols, a joint venture between GreenePublishing Associates, Inc. and John Wiley & Sons, Inc., (supplementedthrough 2005) and PCR Protocols A Guide to Methods and Applications, M.A. Innis et al., eds., Academic Press Inc. San Diego, Calif. (1990).

The PCR process phases of denaturing, annealing, and extension occur atdifferent temperatures and cause target DNA molecule samples toreplicate themselves. Temperature cycling (thermocyling) requirementsvary with particular nucleic acid samples and assays. In the denaturingphase, a double stranded DNA (dsDNA) is thermally separated into singlestranded DNA (ssDNA). During the annealing phase, primers are attachedto the single stand DNA molecules. Single strand DNA molecules grow todouble stranded DNA again in the extension phase through specificbindings between nucleotides in the PCR solution and the single strandDNA. Typical temperatures are 95° C. for denaturing, 55° C. forannealing, and 72° C. for extension. The temperature is held at eachphase for a certain amount of time which may be a fraction of a secondup to a few tens of seconds. The DNA is doubled at each cycle; itgenerally takes 20 to 40 cycles to produce enough DNA for theapplications. To have good yield of target product, one has toaccurately control the sample temperatures at the different phases to aspecified degree.

More recently, a number of high throughput approaches to performing PCRand other amplification reactions have been developed, for example,involving amplification reactions in microfluidic devices, as well asmethods for detecting and analyzing amplified nucleic acids in or on thedevices. Thermal cycling of the sample for amplification is usuallyaccomplished in one of two methods. In the first method, the samplesolution is loaded into the device and the temperature is cycled intime, much like a conventional PCR instrument. In the second method, thesample solution is pumped continuously through spatially varyingtemperature zones. See, for example, Lagally et al. (AnalyticalChemistry 73:565-570 (2001)), Kopp et al. (Science 280:1046-1048(1998)), Park et al. (Analytical Chemistry 75:6029-6033 (2003)), Hahn etal. (WO 2005/075683), Enzelberger et al. (U.S. Pat. No. 6,960,437) andKnapp et al. (U.S. Patent Application Publication No. 2005/0042639).

Many detection methods require a large number of copies (millions, forexample) of the original DNA molecule, in order for the DNA to becharacterized. Because the total number of cycles is fixed with respectto the number of desired copies, the only way to reduce the process timeis to reduce the length of a cycle. Thus, the total process time may besignificantly reduced by rapidly heating and cooling samples to processphase temperatures while accurately maintaining those temperatures forthe process phase duration.

Accordingly, what is desired is a system and method for rapidly andaccurately changing process temperatures in PCR and thermal meltprocesses.

SUMMARY

In one aspect, the present invention provides an improved tunabletemperature measurement circuit. In some embodiments, the improvedtunable temperature measurement circuit includes a source nodemaintained at a predetermined source voltage; a ground node maintainedat a predetermined ground voltage; and a bridge circuit coupled toprogrammable gain instrumentation amplifier. In one embodiment, thebridge circuit comprises (1) a first resistance temperature detectorconnected between the source node and a first measurement node, (2) afirst reference resistor connected between the first measurement nodeand the ground node, (3) a potentiometer (e.g. a programmable digitalpotentiometer) connected between the source node and a reference node,and (4) a scaling resistor connected between the reference node and theground node. The programmable gain instrumentation amplifier may beconnected so that a first input to the first programmable gaininstrumentation amplifier is connected to the reference node, a secondinput to the first programmable gain instrumentation amplifier isconnected to the first measurement node, and the output of the firstprogrammable gain instrumentation amplifier is representative of thetemperature sensed by the first resistance temperature detector. In someembodiments of the improved tunable temperature measurement circuit, oneor more of the first reference resistor and the scaling resistor arealso potentiometers.

In some embodiments, the improved tunable temperature measurementcircuit also includes a capacitor connected in parallel with the scalingresistor and/or a low-pass filter coupled to the output of the firstprogrammable gain instrumentation amplifier.

In some embodiments, the improved tunable temperature measurementcircuit also includes a bypass circuit connected between the firstmeasurement node and the ground node, wherein the bypass circuitcomprises a bypass switch (e.g., a digital switch) in series with abypass resistor. In some embodiments, the bypass circuit is configuredto pulse width modulate a current passing through the first resistancetemperature detector.

In some embodiments, the improved tunable temperature measurementcircuit also includes a power control circuit connected to the firstmeasurement node, wherein the power control circuit comprises a bottompower switch connected between the measurement node and a bottom powernode maintained at the predetermined source voltage, and a groundingswitch connected in series with a bypass resistor between themeasurement node and the ground node. In some embodiments, the tunabletemperature measurement circuit may also include a shunt circuitconnected between the reference resistor and the ground node, whereinthe shunt circuit comprises a shunt switch in parallel with a shuntresistor.

In some embodiments, the improved tunable temperature measurementcircuit also includes: a selector switch disposed in between the firstresistance temperature detector and the first measurement node; and oneor more second resistance temperature detectors connected to the sourcenode in parallel with the first resistance temperature detector. Inthese embodiments, the selector switch may be configured to connect oneof the first resistance temperature detector and the one or more secondresistance temperature detectors to the measurement node.

In some embodiments, the improved tunable temperature measurementcircuit also includes: a second resistance temperature detectorconnected between the source node and a second measurement node, asecond reference resistor connected between the second measurement nodeand the ground; and a second programmable gain instrumentationamplifier. In these embodiments, a first input to the secondprogrammable gain instrumentation amplifier is connected to thereference node, a second input to the second programmable gaininstrumentation amplifier is connected to the second measurement node,and the output of the second programmable gain instrumentation amplifieris representative of the temperature sensed by the second resistancetemperature detector. In some embodiments, the improved tunabletemperature measurement circuit also includes a unity gain buffer,wherein the reference node is connected to the programmable gaininstrumentation amplifiers via the unity gain buffer.

In another aspect, the invention provides a method of calibrating thepotentiometer in an improved tunable temperature measurement systemincluding the improved tunable temperature measurement circuit. In someembodiments, the method of calibrating the potentiometer includes thesteps of: (a) setting the resistance value of the potentiometer to afirst resistance value; (b) setting the gain of the first programmablegain instrumentation amplifier to a first gain value; (c) measuring thevoltage output from the first programmable gain instrumentationamplifier; (d) in the case that the measured voltage is above apredetermined target value (e.g., a value selected to maximize thesignal to noise ratio in the output of the first programmable gaininstrumentation amplifier), adjusting the resistance value of thepotentiometer in a first direction; (e) in the case that the measuredvoltage is below the predetermined target value, adjusting theresistance value of the potentiometer in a direction opposite to thefirst direction; and (f) repeating steps (c) through (e) until themeasured voltage from the first programmable gain instrumentationamplifier is equal to the predetermined target value.

In some embodiments, the method of calibrating the potentiometer in animproved tunable temperature measurement system of also includes thesteps of: (g) after performing step (f), storing the resistance value ofthe potentiometer in an electronic memory; (h) associating the storedresistance value with an identifier corresponding to the firstresistance temperature detector; (i) repeating steps (a) through (h) fora plurality of resistance temperature detectors to create a plurality ofassociations between resistance temperature detectors and resistancevalues; (j) detecting the presence of one of the plurality of resistancetemperature detectors; and (k) setting the resistance value of thepotentiometer to the resistance value associated with the one of theplurality of resistance temperature detectors. In some embodiments, thestep of detecting the presence of one of the plurality of resistancetemperature detectors comprises reading a machine readable bar code oran RFID tag from a platform chip containing the one of the plurality ofresistance temperature detectors.

In another aspect, the invention provides a method of calibrating theself-heating properties of the improved tunable temperature measurementsystem. In some embodiments, the method of calibrating the self-heatingproperties includes: (a) setting the predetermined source voltage to afirst source voltage value corresponding to a desired operational supplyvoltage; (b) setting the gain of the first programmable gaininstrumentation amplifier to a first gain value corresponding to adesired operational gain value; (c) measuring the voltage output fromthe first programmable gain instrumentation amplifier; (d) determining afirst ratio of the output from the first programmable gaininstrumentation amplifier to the source node voltage multiplied by thegain of the first programmable gain instrumentation amplifier; (e)decreasing the predetermined source voltage to a new source voltagevalue; (f) measuring the voltage output from the first programmable gaininstrumentation amplifier; (g) determining a new ratio of the outputfrom the first programmable gain instrumentation amplifier to themeasured source node voltage multiplied by the gain of the firstprogrammable gain instrumentation amplifier; (h) determining anasymptote ratio by repeating steps (e) through (g) until the change ofthe new ratio determined at (g) between subsequent iterations is beneatha predetermined threshold; and (i) determining an operationalself-heating voltage difference by multiplying the desired operationalgain value by the difference between the first ratio and the asymptoteratio.

In some embodiments of the method of calibrating the self-heatingproperties of the improved tunable temperature measurement system, steps(c) and (f) further comprise measuring the voltage at the source node;and steps (d) and (g) use the measured voltage at the source node as thesource node voltage.

In some embodiments of the method of calibrating the self-heatingproperties of the improved tunable temperature measurement system, step(e) further comprises increasing the gain of the first programmable gaininstrumentation amplifier to a new gain value such that the product ofthe first source voltage value and the first gain value is equal to theproduct of the new source voltage value and the new gain value.

In another aspect, the invention provides a method for performingthermal calibration of the improved tunable temperature measurementsystem comprising the steps of: (a) setting the predetermined sourcevoltage to a desired operational supply voltage; (b) setting the gain ofthe first programmable gain instrumentation amplifier to a desiredoperational gain value; (c) bringing the resistance temperature detectorto a known temperature (e.g. by utilizing an externally controlledheating device that has been independently calibrated such as a Peltierdevice or a resistive heater); (d) measuring a voltage output from thefirst programmable gain instrumentation amplifier; (e) storing themeasured output voltage in an electronic memory in association with theknown temperature; (f) repeating steps (c) through (e) to store aplurality of associations between known temperatures and correspondingmeasured output voltages; and (g) utilizing the stored associations tocalibrate the circuit for thermal variations (e.g. by utilizing a lookup table for the plurality of known temperatures or by calculating asuitable curve to interpolate output voltage between the knowntemperatures).

In another aspect, the invention provides a system of controlling thetemperature of a microfluidic device for performing biologicalreactions. In some embodiments, the system of controlling thetemperature of a microfluidic device for performing biological reactionsincludes an improved tunable temperature measurement circuit comprisinga source node maintained at a predetermined source voltage; a groundnode maintained at a predetermined ground voltage; and a bridge circuitcoupled to programmable gain instrumentation amplifier. The bridgecircuit comprises (1) a first resistance temperature detector connectedbetween the source node and a first measurement node, (2) a firstreference resistor connected between the first measurement node and theground node, (3) a potentiometer connected between the source node and areference node, and (4) a scaling resistor connected between thereference node and the ground node. The programmable gaininstrumentation amplifier may be connected so that a first input to thefirst programmable gain instrumentation amplifier is connected to thereference node, a second input to the first programmable gaininstrumentation amplifier is connected to the first measurement node,and the output of the first programmable gain instrumentation amplifieris representative of the temperature sensed by the first resistancetemperature detector.

In some embodiments, the system of controlling the temperature of amicrofluidic device for performing biological reactions includes abridge adjustment controller configured to: (a) set the resistance valueof the potentiometer to a first resistance value; (b) set the gain ofthe first programmable gain instrumentation amplifier to a first gainvalue; (c) measure the voltage output from the first programmable gaininstrumentation amplifier; (d) in the case that the measured voltage isabove a predetermined target value, adjust the resistance value of thepotentiometer in a first direction; (e) in the case that the measuredvoltage is below the predetermined target value, adjust the resistancevalue of the potentiometer in a direction opposite to the firstdirection; and (f) repeat steps (c) through (e) until the measuredvoltage from the first programmable gain instrumentation amplifier isequal to the predetermined target value.

In some embodiments, the system of controlling the temperature of amicrofluidic device for performing biological reactions includes aself-heating calibration controller configured to: (a) set thepredetermined source voltage to a first source voltage valuecorresponding to a desired operational supply voltage; (b) set the gainof the first programmable gain instrumentation amplifier to a first gainvalue corresponding to a desired operational gain value; (c) measure thevoltage output from the first programmable gain instrumentationamplifier; (d) determine a first ratio of the output from the firstprogrammable gain instrumentation amplifier to the source node voltagemultiplied by the gain of the first programmable gain instrumentationamplifier; (e) decrease the predetermined source voltage to a new sourcevoltage value; (f) measure the voltage output from the firstprogrammable gain instrumentation amplifier; (g) determine a new ratioof the output from the first programmable gain instrumentation amplifierto the measured source node voltage multiplied by the gain of the firstprogrammable gain instrumentation amplifier; (h) determine an asymptoteratio by repeating steps (e) through (g) until the change of the newratio determined at (g) between subsequent iterations is beneath apredetermined threshold; and (i) determine an operational self-heatingvoltage difference by multiplying the desired operational gain value bythe difference between the first ratio and the asymptote ratio.

In some embodiments, the system of controlling the temperature of amicrofluidic device for performing biological reactions includes athermal calibration controller configured to: (a) set the predeterminedsource voltage to a desired operational supply voltage; (b) set the gainof the first programmable gain instrumentation amplifier to a desiredoperational gain value; (c) bring the resistance temperature detector toa known temperature; (d) measure a voltage output from the firstprogrammable gain instrumentation amplifier; (e) store the measuredoutput voltage in an electronic memory in association with the knowntemperature; (f) repeat steps (c) through (e) to store a plurality ofassociations between known temperatures and corresponding measuredoutput voltages; and (g) utilize the stored associations to calibratethe circuit for thermal variations

The above and other embodiments of the present invention are describedbelow with reference to the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are incorporated herein and form partof the specification, illustrate various embodiments of the presentinvention. In the drawings, like reference numbers indicate identical orfunctionally similar elements. Additionally, the left-most digit(s) of areference number identifies the drawing in which the reference numberfirst appears.

FIG. 1 is a block diagram of a system for performing PCR and thermalmelt analysis.

FIG. 2 is a diagram of a microfluidic chip.

FIG. 3 a is a circuit diagram illustrating an improved tunabletemperature measurement circuit according to an embodiment of theinvention.

FIG. 3 b is a circuit diagram illustrating a programmable gaininstrumentation amplifier according to an embodiment of the invention.

FIG. 3 c is a circuit diagram illustrating a capacitor connected inparallel with the scaling resistor according to some embodiments of theinvention.

FIG. 3 d is a circuit diagram illustrating a low pass filter applied tothe output of the programmable gain instrumentation amplifier accordingto some embodiments of the invention.

FIG. 4 is a circuit diagram illustrating a bypass circuit according toan embodiment of the invention.

FIG. 5 is a circuit diagram illustrating a selector switch according toan embodiment of the invention.

FIG. 6 a is a circuit diagram illustrating an improved tunabletemperature measurement circuit according to an embodiment of theinvention.

FIG. 6 b is a circuit diagram illustrating a sensing portion of animproved tunable temperature measurement circuit with a plurality ofmeasurement nodes according to an embodiment of the invention.

FIG. 6 c is a circuit diagram illustrating an amplification andmeasurement portion of an improved tunable temperature measurementcircuit with a plurality of measurement nodes according to an embodimentof the invention.

FIG. 6 d is a circuit diagram illustrating a programmable gaininstrumentation amplifier according to an embodiment of the invention.

FIG. 6 e is a circuit diagram illustrating a power supply circuit for atunable temperature measurement circuit with a plurality of programmablegain instrumentation amplifiers according to an embodiment of theinvention.

FIG. 6 f is a circuit diagram illustrating control circuitry for abypass circuit that performs multiplex measurements according to anembodiment of the invention.

FIG. 7 a is a circuit diagram illustrating an improved tunabletemperature measurement circuit with a unity gain buffer according to anembodiment of the invention.

FIG. 7 b is a circuit diagram illustrating a unity gain buffer accordingto an embodiment of the invention.

FIG. 8 is a circuit diagram illustrating an improved tunable temperaturemeasurement circuit according to an embodiment of the invention.

FIG. 9 is a flow chart illustrating a bridge adjustment method inaccordance with embodiments of the invention.

FIG. 10 a is a flow chart illustrating a self-heating calibration methodin accordance with embodiments of the present invention.

FIG. 10 b is a flow chart illustrating a self-heating calibration methodin accordance with embodiments of the present invention.

FIG. 10 c is a flow chart illustrating a self-heating calibration methodin accordance with embodiments of the present invention.

FIG. 11 is a graph illustrating the ratio V_(out)/G(V_(cc)) for varyingvalues of V_(cc) in a tunable temperature measurement circuit inaccordance with an embodiment of the present invention.

FIG. 12 is a flow chart depicting a thermal calibration method inaccordance with embodiments of the present invention.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

FIG. 1 illustrates a functional block diagram of a system 100 forperforming PCR and thermal melt analysis according to some embodimentsof the invention. As illustrated in FIG. 1, system 100 may include amicrofluidic device 102. Microfluidic device 102 may include one or moremicrofluidic channels 104. In the examples shown, device 102 includestwo microfluidic channels, channel 104 a and channel 104 b. Althoughonly two channels are shown in the exemplary embodiment, it iscontemplated that device 102 may have fewer than two or more than twochannels. For example, in some embodiments, device 102 includes eightchannels 104.

Device 102 may include two DNA processing zones, a DNA amplificationzone 131, sometimes referred to herein as PCR zone 131, and a DNAmelting zone 132. A DNA sample traveling through the PCR zone 131 mayundergo PCR, and a DNA sample passing through melt zone 132 may undergohigh resolution thermal melting. As illustrated in FIG. 1, PCR zone 131includes a first portion of channels 104 and melt zone 132 includes asecond portion of channels 104, which is down stream from the firstportion.

In order to achieve PCR for a DNA sample flowing through the PCR zone131, the temperature of the sample must be cycled, as is well known inthe art. Accordingly, in some embodiments, system 100 includes atemperature control system 120. The temperature control system 120 mayinclude a temperature sensor, a heater/cooler, and a temperaturecontroller. In some embodiments, a temperature control system 120 isinterfaced with a main controller 130 so that main controller 130 cancontrol the temperature of the samples flowing through the PCR zone andthe melting zone.

Main controller 130 may be connected to a display device for displayinga graphical user interface. Main controller 130 may also be connected touser input devices which allow a user to input data and commands intomain controller 130.

To monitor the PCR process and the melting process that occur in PCRzone 131 and melt zone 132, respectively, system 100 may include animaging system 118. Imaging system 118 may include an excitation source,an image capturing device, a controller, and an image storage unit.Other aspects of a suitable system in accordance with some aspects ofthe invention are disclosed in U.S. patent application Ser. No.11/770,869, incorporated herein by reference in its entirety.

FIG. 2 illustrates another embodiment of a microfluidic device 202. Asshown in FIG.2, the microfluidic chip 202 may comprise a plurality ofmicrofluidic channels 204 a-204 h that traverse heater/sensor regions206, 208. The plurality of microfluidic channels 204 a-204 h may bedisposed in alignment with heaters/sensors in a first region 206configured to perform PCR and may also be disposed in alignment withheaters/sensors in a second region 208 configured to perform highresolution thermal melting. In some embodiments, heaters/sensors in theheater/sensor regions 206, 208 comprise resistance temperature detectors(“RTD”s). The RTDs may be composed from a variety of materials, forexample, the RTDs may be platinum, copper, and/or nickel RTDs.

The temperature-dependent resistance Rh of an RTD may be related to itstemperature by the linear approximation shown in Equation 1:

Rh(T)=Rh(T ₀)·[1+α(T−T ₀)]  Equation 1

In equation 1, Rh(T₀) is a known resistance of the RTD at apredetermined temperature T₀, T is the current temperature of the RTD,and a is a linear temperature coefficient of resistance.

As the temperature T of the RTD increases, its resistance Rh will alsoincrease according to Equation 1. Therefore, an accurate measurement ofRh(T) can indicate the temperature T of the RTD as shown in Equation 2below:

$\begin{matrix}{T = {T_{0} + \frac{\frac{{Rh}(T)}{{Rh}\left( T_{0} \right)} - 1}{\alpha}}} & {{Equation}\mspace{14mu} 2}\end{matrix}$

One method of measuring the resistance Rh of the RTD is with a simplevoltage dividing circuit. In some embodiments of a voltage dividingcircuit, the RTD is placed in series with a reference resistor having aresistance Ri. The reference resistor may comprise a single resistiveelement, or in some embodiments may comprise a plurality of discreteelements that, in combination, exhibit a relatively constant resistancevalue. The RTD in series with the reference resistor are connectedbetween a known voltage difference (e.g., V_(cc)−V_(G)), and a voltagemeasurement V_(i) is taken at a point between the RTD and the referenceresistor. In this configuration, the resistance Rh of the RTD can becalculated with the following relations:

$\begin{matrix}{\left( {V_{i} - V_{G}} \right) = {\left( {V_{cc} - V_{G}} \right) \cdot \frac{Ri}{{{Rh}(T)} + {Ri}}}} & {{Equation}\mspace{14mu} 3} \\{{{Rh}(T)} = {{Ri} \cdot \left\lbrack {\frac{\left( {V_{cc} - V_{G}} \right)}{\left( {V_{i} - V_{G}} \right)} - 1} \right\rbrack}} & {{Equation}\mspace{14mu} 4}\end{matrix}$

By substituting the expression for Rh from equation 4 into equation 2,one can determine the temperature of the RTD by measuring the voltageV_(i):

$\begin{matrix}{T = {T_{0} + \frac{{\frac{Ri}{{Rh}\left( T_{0} \right)} \cdot \left\lbrack {\frac{\left( {V_{cc} - V_{G}} \right)}{\left( {V_{i} - V_{G}} \right)} - 1} \right\rbrack} - 1}{\alpha}}} & {{Equation}\mspace{14mu} 5}\end{matrix}$

Accordingly, one can accurately determine the temperature T of the RTDwhen Rh(T₀), α, V_(cc), and V_(G) are known.

It is desirable to maximize the sensitivity of the measurement tochanges in temperature (i.e., maximize ΔV_(i)/ΔT). The sensitivity ismaximized when the resistance of the RTD is equal to the resistance ofthe reference resistor. Accordingly, the nominal value for theresistance Ri of the reference resistor preferably should be selected tobe approximately equal to the resistance Rh of the RTD over the expectedrange of temperatures in which the measurement circuit will be used.

To better utilize the voltage range of the measurement V_(i) and toimprove the signal to noise ratio of the temperature detection, inaccordance with one embodiment the RTD may be placed within a bridgecircuit. FIG. 3 a illustrates a circuit diagram for a tunabletemperature measurement circuit 300 in accordance with one embodiment ofthe present invention. As shown in FIG. 3 a, the tunable temperaturemeasurement circuit 300 may comprise a source node 303 maintained at apredetermined source voltage (V_(cc)), a ground node 304 maintained at apredetermined ground voltage (V_(G)), and a bridge circuit 301 coupledto a differential amplifier 302, which in some embodiments may be aprogrammable gain instrumentation amplifier. Differential amplifier 302is referred to herein as a programmable gain instrumentation amplifier,but the differential amplifier in some embodiments need not be limitedto a programmable gain instrumentation amplifier. In one embodiment, thebridge circuit 301 may comprise a first leg (“measurement leg”) 301 aincluding an RTD 305 having a temperature-dependent resistance Rh(T)connected between the source node 303 and a first measurement node 306and a reference resistor 307 having a resistance Ri connected betweenthe first measurement node 306 and the ground node 304. The bridgecircuit may also comprise a second leg (“reference leg”) 301 b includinga potentiometer 308 having an adjustable resistance Rj connected betweenthe source node 303 and a reference node 309, and a scaling resistor 310having a resistance Rk connected between the reference node 309 and theground node 304. In some embodiments, potentiometer 308 is aprogrammable digital potentiometer.

In one exemplary embodiment, the relative resistances in the bridgecircuit 301 are selected to be proportional so that:

Rj=C·Rh(T ₀)   Equation 6

Rk=C·Ri   Equation 7

In the above relations, C is a scaling factor that ensures the currentthrough the second leg is small, which in turn ensures that thepotentiometer 308 and the scaling resistor 310 do not heatsignificantly. In one embodiment, this scaling factor may be in a rangebetween 1 and 1000, and is preferably 100.

In temperature measurement circuit 300, the voltage difference betweenmeasurement node 306 (V_(i)) and reference node 309 (V_(k)) are comparedvia instrumentation amplifier 302 to output a precise signalcorresponding to the temperature sensed by RTD 305.

FIG. 3 b illustrates one embodiment of a programmable gaininstrumentation amplifier 302. As shown in FIG. 3 b, the relationshipbetween the voltage out V and the difference between the input voltagesV_(i) and V_(k) can be controlled by adjusting potentiometer R1. In someembodiments, potentiometer R1 is a programmable digital potentiometer.

In the embodiment shown in FIG. 3 a, the voltage V_(i) of themeasurement node 306 is coupled to the non-inverting input of theprogrammable gain instrumentation amplifier 302, while the voltage V_(k)from the reference node 309 is coupled to the inverting input of theprogrammable gain instrumentation amplifier 302. In alternateembodiments, the voltage V_(i) of the measurement node 306 may becoupled to the inverting input of the programmable gain instrumentationamplifier 302, while the voltage V_(k) from the reference node 309 iscoupled to the non-inverting input of the programmable gaininstrumentation amplifier 302.

Referring to FIGS. 3 c and 3 d, certain features contained in someembodiments of a temperature measurement circuit 300 are illustrated. Asshown in FIG. 3 c, the reference leg 301 b of the bridge circuit 301 mayalso comprise a capacitor 311 connected in parallel with the scalingresistor 310. This capacitor 311 tends to make the reference side of thebridge circuit 301 more stable and less prone to high frequency noise.As shown in FIG. 3 d, a low pass filter may be coupled to the output ofthe instrumentation amplifier 302. The filter may comprise a resistor312 and a capacitor 313. The resistance of the resistor 312 and thecapacitance of the capacitor 313 are selected to filter out highfrequency noise such as, for example, variations at a rate exceedingapproximately 10 kHz. In some embodiments the resistance of the resistor312 may be in a range between 200Ω and 10,000Ω, and is preferably 820Ω.In other embodiments, the capacitance of the capacitor 313 may be in arange between 100 pF and 40,000 pF, and is preferably 500 pF.

FIG. 4 illustrates a tunable temperature measurement circuit 300 inaccordance with other embodiments. As shown in FIG. 4, the temperaturemeasurement circuit 300 may further comprise a bypass circuit 400 forproviding greater control over the amount of current passing through theRTD 305. The bypass circuit 400 may include a switch 410, such as adigital switch, and a low resistance resistor 411 having resistance Rd.The low resistance resistor 411 is switched into the circuit to maximizethe current flowing through the RTD 305. In some embodiments theresistance of the low resistance resistor 411 may be in a range between0Ω and 1,000Ω, and is preferably 0Ω. In other embodiments, theresistance of the low resistance resistor 411 is selected to besubstantially smaller than the resistance Ri of the reference resistor307. In some embodiments, switch 410 is used in conjunction with pulsewidth modulation to enable greater control over the current through RTD305.

Referring to FIG. 5, another embodiment of the tunable temperaturemeasurement circuit 300 is illustrated. As shown in FIG. 5, thetemperature measurement circuit may be configured to accommodate aplurality of RTDs (e.g. RTDs 505 a and 505 b) using switching circuitry.The plurality of RTDs may be configured to measure the temperatures ofdifferent environments or may be selected to possess differenttemperature-dependent resistances appropriate for measuring differentranges of temperatures. In some embodiments, the switching circuitry maycomprise a selector switch 512. As shown in FIG. 5, the selector switch512 can be configured to connect one of the RTDs 505 a, 505 b to themeasurement node 306. The potentiometer 308 and programmable gaininstrumentation amplifier 302 may be adjusted to accommodate each RTD asit is switched into the circuit.

FIG. 6 a illustrates yet another embodiment of temperature detectorcircuit 300. As shown in FIG. 6 a, the bridge circuit may be configuredto simultaneously measure the resistance of a plurality of RTDs byproviding one or more additional measurement legs 601. In someembodiments, the additional measurement leg 601 may comprise anadditional RTD 605 and an additional reference resistor 607.Additionally, the embodiment shown in FIG. 6 a includes an additionalprogrammable gain instrumentation amplifier 602 coupled to a measurementnode 606 of the additional measurement leg 601. In this embodiment,reference node 309 may be coupled to the plurality of the programmablegain instrumentation amplifiers 302, 602 and thus a single referencenode 309 may provide a reference voltage for the plurality of RTDs 305,605.

Referring to FIG. 6 b, a relationship between a microfluidic chip 102,202 and the measurement legs 601 of the temperature detecting circuit300 according to one embodiment is illustrated. As shown in FIG. 6 b, aplurality of measurement legs 601 (e.g. 601 a through 601 h) may beincorporated into a microfluidic chip 102, 202 while the remainingportion of the temperature detecting circuit is incorporated into atemperature control system, e.g. temperature control system 120. Thisconfiguration can enable a single temperature measurement circuit to beused with a plurality of distinct microfluidic chips and can alsosimplify the fabrication of the microfluidic chips by reducing thecircuitry thereon.

FIG. 6 c illustrates an exemplary embodiment of the programmable gaininstrumentation amplifiers coupled to the measurement legs 601 a though601 h shown in FIG. 6 b. As shown in FIG. 6 c, the measurement node(AI-0 through AI-7) for each measurement leg 601 a-601 h is connected toa separate programmable gain instrumentation amplifier 602 a through 602h. Furthermore, as shown in FIG. 6 c, the voltage V_(k) of the referencenode 309 is shared among all of the programmable gain instrumentationamplifiers in accordance with one embodiment. As described withreference to FIG. 6 b, in some embodiments the programmable gaininstrumentation amplifiers 602 a through 602 h and associated controlcircuitry may be incorporated into a temperature control system 120rather than integrated with the microfluidic chip 102, 202. Asillustrated in FIG. 6 d, in some embodiments, the programmable gaininstrumentation amplifiers may be implemented using, for example, SingleResistor Gain Programmable, Precision Instrumentation Amplifiers, LinearTechnology part no. LT1167.

Referring to FIGS. 6 e and 6 f, schematics of circuitry for performing amultiplex measurement technique according to some embodiments of theinvention are illustrated. In one embodiment, the multiplex measurementtechnique involves switching a common power supply circuit 650 and aplurality of switching circuits 651 (see FIG. 6 f) to operate each RTDmeasurement leg (e.g. 601 a through 601 h) independently.

FIG. 6 e illustrates the common power supply circuit 650. Power supplycircuit 650 may comprise electric switches 652, which can be Metal OxideSemiconductor Field Effect Transistors (MOSFET) switches, that aredriven by a digital line “Top Power.” The switches 652 may be used toconnect or disconnect a top power source 653 (e.g. +30V) to a commonlead 654.

FIG. 6 f illustrates one embodiment of the switching circuit 651 for oneof the eight RTD measurement legs (e.g., the measurement leg 601 acorresponding to RTD 605 a). The switching circuits 651 may includeelectric switches 655, which can be MOSFET switches, that are driven bya digital line “Bottom Power,” for connecting the corresponding RTD(e.g. RTD 605 a) with a bottom power source 656 (e.g. +30V).

The switching circuits 651 may also include electric switch 657, whichcan be a MOSFET switch, that is driven by a digital line “Heater,” forconnecting the corresponding RTD (e.g. 605 a) with a reference voltage658 (e.g. 0 V) via a low resistance resistor 659 having a relatively alow resistance (e.g. 0Ω). In some embodiments, the resistance of the lowresistance resistor 659 may be in a range between 0Ω and 1,000Ω, and ispreferably 0Ω. The low resistance resistor 659 is switched into thecircuit to maximize the current flowing through the RTD 605 a and causethe RTD 605 a to heat rapidly.

The switching circuits 651 may also include electric switch 660, whichcan be a MOSFET switch, that is driven by a digital line “Measure,” forconnecting the corresponding RTD (e.g. 605 a) with the reference voltage658 via a reference resistor 607 a having a relatively high resistance(e.g. 1,000Ω). In some embodiments, the resistance of the low resistanceresistor 659 is selected to be substantially smaller than the resistanceof the reference resistor 607 a.

Additionally, as illustrated in FIG. 6 f, in some embodiments, theswitching circuits 651 may include a shunt resistor 661 connected inparallel with the switch 660 and having a resistance value substantiallyhigher than the reference resistor 607 a (e.g. 1,000,000Ω). The shuntresistor 661 acts as a shunt around switch 660 when switch 660 is OFF.With the switch 660 ON, the resistance measurements can be taken asnormal. When the switch 660 is OFF, however, then resistancemeasurements can still be taken due to the small current that stillflows through the shunt resistor 661.

In one embodiment, each of the remaining RTD measurement legs (e.g. 601b-601 h) also includes a switching circuit 651. With the circuitillustrated in FIGS. 6 e and 6 f, the common lead 654 can bedisconnected from the top power source 653, each RTD can be selectivelyconnected to the bottom power source 656 and reference voltage 658, andeach RTD can be selectively removed from the resistive network. Thisembodiment thus allows for isolated, power-on and power-offmeasurements.

In some embodiments, the circuit illustrated in FIGS. 6 e and 6 f may beused to measure the series resistance across any two of the RTDs (e.g.,605 a-605 h). For example, the Top Power signal may be used to set theswitches 652 to the OFF state and disconnect the top power source 653from the common lead 654. Next, the Bottom Power signal corresponding toa first switching circuit 651 may be used to connect the bottom powersource 656 to the corresponding first RTD (e.g., 605 a). Then, theMeasure signal of a second switching circuit 651 may be used to connectthe reference voltage 658 to the corresponding second RTD (e.g., 605 b)via the corresponding reference resistor (e.g. 607 b). These settingswill cause current to flow from the bottom power source 656, through thefirst RTD (e.g., 605 a), the second RTD (e.g., 605 b), and the referenceresistor (e.g. 607 b) in series to the reference voltage 658, and thevoltage at the measurement node (e.g. AI-1) of the second switchingcircuit 651 will correspond to the series resistance of the first andsecond RTDs (e.g. 605 a and 605 b). By controlling the switches of eachof the switching circuits 651, many other combinations of resistancescan be measured. Aspects of additional circuitry for performingmultiplex measurement techniques are disclosed in commonly assigned U.S.patent application Ser. No. 12/165,043, incorporated herein by referencein its entirety.

In some embodiments wherein multiple RTDs share a common reference node,the multiplex measurement technique described above can createundesirable cross-talk. In some embodiments, the RTD leads fluctuatebetween three different voltage levels: V_(cc), V_(G) ground, andV_(measure). These three states occur when the RTD is used as the powersupply side for multiplex measurement, when it is heated, and when it isused as a sensor, respectively. Using the multiplex circuit technique,only one channel will be at V_(measure) at a given moment and the otherchannels will be at a voltage close to V_(cc). However, whenever thevoltage of a channel goes to V_(cc) or ground, this can force theinstrumentation amplifier associated with that particular sensor intosaturation (also called the overload condition). Using the sharedreference node configuration, this overload of one instrumentationamplifier may affect the reference voltage V_(k), causing one sensor'soverload to modify another sensor's reference voltage.

FIG. 7 a illustrates an embodiment of the temperature detecting circuit300 that addresses this cross-talk issue. As shown in FIG. 7 a, a unitygain buffer 720 is placed in between the reference node 309 and theprogrammable gain instrumentation amplifiers 302, 602. This bufferreduces the effect of cross-talk between the sensors by preventing theoverload condition of one instrumentation amplifier from affecting thereference voltage V_(k).

FIG. 7 b illustrates a unit gain buffer 720 in accordance with oneembodiment. As shown in FIG. 7 b, a unity gain buffer receives inputvoltage V_(k) from the reference node and outputs voltage V_(k)′ to theprogrammable gain instrumentation amplifiers. The low output impedanceof the unit gain buffer 720 provides the current required by anysaturated instrumentation amplifiers and allows the unsaturatedamplifiers to work as designed without affecting the reference voltage.

Referring to FIG. 8, another embodiment of the temperature detectingcircuit 300 is illustrated. As shown in FIG. 8, in this embodiment, thebridge circuit 301 is fully adjustable with programmable resistances. Inaddition to potentiometer 308, the reference resistor 807 and scalingresistor 810 may also comprise potentiometers. In some embodiments, thepotentiometers may be programmable digital potentiometers. An advantageof this configuration is its ability to accept widely varyingheater/sensor Rh resistances while optimizing the signal to noise ratio.With this configuration it is possible for a single temperaturecontroller 120 to accept different platform chips 102, 202 that performdifferent biological assays. Some assays may require a different PCRprotocol with additional thermocycling (e.g. small amplicon or probemelting). The added thermocycling may be achieved with a longermicrochannel and correspondingly longer thin-film heater/sensor Rh. Thelonger heater/sensor Rh would then likely have a significantly largerresistance. By adding programmable reference resistor 807, improvedsensitivity can be maintained, such as, for example, when Rh=Ri. In someembodiments, programmable resistor 810 is included to maintain a fixedbridge ratio C even while the resistance Ri of the reference resistor807 varies.

In one exemplary embodiment, if the RTDs in platform chip “A” haveresistances of 100Ω and the RTDs in platform chip “B” have resistancesof 250Ω, then the programmable reference resistor 807 could be digitallyadjusted to 100Ω or 250Ω as required. Of course, there is no limit tothe number of different platform chips that could be used because thereference resistor 807 and the scaling resistor 810 are programmed asrequired. Furthermore, a single platform chip could have two or morevery different RTD resistances. This may improve the functionality ofthe device by enabling two or more very different kinds of assays on thesame chip (i.e. a hybrid chip that simultaneously runs two verydifferent PCR protocols). In this case, the resistors 807 and 810 wouldsimply be programmed different for each RTD.

Finally, this fully adjustable bridge configuration could be used alongwith the shared bridge configuration discussed above to create a highlyflexible, but still efficient, measurement system.

Bridge Adjustment

In some embodiments, the temperature-dependent resistance Rh(T) of theRTD may vary due to manufacturing variations, contact irregularities,corrosion, differences in design, etc. In a bridge adjustment processthe potentiometer 308 in the tunable temperature measurement circuit 300is tuned to account for variable RTD characteristics.

FIG. 9 illustrates a flow chart describing a bridge adjustment process900 for adjusting the potentiometer 308 in the temperature measurementcircuit 300 in accordance with another aspect of the present invention.In some embodiments, the bridge adjustment process 900 may be performedby a bridge adjustment controller. Process 900 may begin at 920, wherethe bridge adjustment controller may set an initial resistance value forthe potentiometer 308. At step 930, the bridge adjustment controller mayset an initial gain value G for the programmable gain instrumentationamplifier 302.

After these initial values are set at steps 920 and 930, the bridgeadjustment controller measures the output voltage V_(out) in step 940.As discussed above, the output voltage will be indicative of the gainvalue G multiplied by the difference between the voltage V_(i) of themeasurement node 306 and the voltage V_(k) of the reference node 309.For example, in embodiments where the measurement node 306 is coupled tothe non-inverting input of the programmable gain instrumentationamplifier 302, and the reference node 309 is coupled to the invertinginput of the programmable gain instrumentation amplifier 302, the outputvoltage V_(out) may follow the relation shown in Equation 8:

V _(out) =G·(V−V _(k))   Equation 8

At step 950, the bridge adjustment controller compares the outputvoltage V_(out) against a predetermined target voltage.

In the case that the output voltage V_(out) is above the target voltage,at step 960 the bridge adjustment controller adjusts the resistance ofthe potentiometer in a first direction. For example, in embodimentswhere the measurement node 306 is coupled to the non-inverting input ofthe programmable gain instrumentation amplifier 302, and the referencenode 309 is coupled to the inverting input of the programmable gaininstrumentation amplifier 302, a voltage V_(out) above the targetvoltage may indicate that the voltage V_(k) at the reference node 309 istoo low and the bridge adjustment controller decreases the resistance ofthe potentiometer 308 in order to increase the voltage V_(k) at thereference node 309.

In the case that the output voltage V_(out) is below the target voltage,at step 970 the bridge adjustment controller adjusts the resistance ofthe potentiometer in a second direction. For example, in embodimentswhere the measurement node 306 is coupled to the non-inverting input ofthe programmable gain instrumentation amplifier 302, and the referencenode 309 is coupled to the inverting input of the programmable gaininstrumentation amplifier 302, a voltage V_(out) below the targetvoltage may indicate that the voltage V_(k) at the reference node 309 istoo high and the bridge adjustment controller increases the resistanceof the potentiometer 308 in order to decrease the voltage V_(k) at thereference node 309.

In the case that the output voltage V_(out) is about equal to the targetvoltage (that is, the voltage difference between the reference node 309and the measurement node 306 is within a predetermined margin), then thebridge adjustment controller may terminate the bridge adjustment process900.

In some preferred embodiments, the predetermined target voltage isselected to utilize more of the range of the programmable gaininstrumentation amplifier 302 and improve the signal to noise ratio. Forexample, a target output voltage of zero (0) volts may be selected tomaximize the signal with respect to the common mode voltage (i.e. tomaximize ΔV_(out)/V_(i)). Common target output voltage ranges include 0to +10 V, −5 to +5 V, and −10 to +10 V. In some preferred embodiments,the target output voltage is at the lowest voltage at the lowesttemperature and the highest voltage at the highest temperature (or viceversa).

In some embodiments, the RTD 305 may be integrated with a platform chip(e.g. the microfluidic device 202), while the potentiometer 308 may beintegrated into a temperature control system (e.g. the temperaturecontrol system 120). In this embodiment, many distinct microfluidicdevices or other temperature-controlled devices may be used with thesame control system 120. Each microfluidic device 202 may be marked witha machine readable identification, e.g., a machine readable bar code ora radio-frequency identification (“RFID”) tag. The temperature controlsystem may read the machine readable identification and store thecalibrated potentiometer setting for each device in association with theidentification for that device. The temperature control system may alsobe configured to detect the machine readable identification of a deviceand program the potentiometer for that device based upon the previouslystored settings.

Self-Heating Calibration

According to another aspect of the present invention, a self-heatingcalibration process is used to account for undesirable self-heatingeffects of the RTD.

Using the RTD 305 as a temperature sensor requires passing an electricalcurrent I through it. According to Joule's first law, an electricalcurrent I passing through a resistor having resistance Rh(T) willdissipate an amount of power P as heat:

P=I ² ·Rh(T)   Equation 9

The heating from P will cause the RTD to rise in temperature by anamount ΔT above the actual environmental temperature. This increase intemperature above the environmental temperature is known as self-heatingand can cause undesirable errors in measurement values. The specificmagnitude of the temperature increase will depend upon the rate at whichheat is being produced and the thermal resistance θ between RTD 305 andthe environment:

ΔT=θ·P   Equation 10

Thus, the RTD 305 itself will be at a temperature ΔT higher than theenvironment. In general, ΔT is independent of the ambient temperature.For example, if ΔT is 5° C. and the environment is 20° C., then thetemperature of RTD 305 will be 25° C. In an otherwise comparableenvironment (that is, in an environment where θ has not changedsignificantly) at 100° C., the temperature of the RTD 305 will be 105°C.

The undesirable effects of self-heating can be minimized by limiting thecurrent through the RTD 305. This can be accomplished by increasing theresistance Ri of the reference resistor 307 or by reducing the supplyvoltage V_(cc). However, during operation, both of these changes couldhave undesirable consequences. As noted above, it is desirable incertain embodiments to match the resistance Ri of the reference resistor307 to the resistance Rh of the RTD 305 in order to increase thesensitivity of the system. While increasing the resistance Ri of thereference resistor 307 would reduce the size of the self-heating effectΔT, it would also reduce the sensitivity of the measurement leg of thebridge circuit 301. Furthermore, a high supply voltage V_(cc) is oftendesired for a good common mode signal as well as for the ability torapidly heat the RTD.

Accordingly, it may be preferable to calibrate temperature measurementsystems by calculating the self-heating voltage change ΔV_(sh) that willoccur on the output voltage V_(out) under normal conditions (e.g., thedesired resistance Ri of the reference resistor 307, the desired gainsetting G for the programmable gain instrumentation amplifier 302, andthe desired supply voltage V_(cc)) and simply remove this known errorfrom measurements.

According to some embodiments, the self-heating calibration process maycomprise identifying voltage settings at which self-heating isminimized; measuring the resistance Rh of the RTD when self-heating isminimized; and comparing that value with the resistance Rh of the RTDunder voltage settings in which self-heating is present. As describedabove and with reference to Equation 1, in the absence of self-heatingan RTD may behave as an ohmic device (i.e., exhibit a linear I-V curve)if the ambient temperature T is held constant. Accordingly, in someembodiments the voltage settings at which self-heating is minimized canbe indirectly determined by identifying the voltage settings at whichthe resistance Rh of the RTD remains relatively constant with respect tochanges in voltage, that is, the voltage settings as whichΔRh/ΔV_(cc)≈0. As explained below, in some embodiments these voltagesettings can also be identified as settings at whichΔV_(out)/(G·ΔV_(cc)) remains constant.

The voltage difference between the measurement node 306 and thereference node 309 (V_(i)−V_(k)) is proportional to the voltagedifference between the source voltage and the ground voltage(V_(cc)−V_(G)), as illustrated in Equation 11:

$\begin{matrix}{\left( {V_{i} - V_{k}} \right) = {\left( {V_{cc} - V_{G}} \right) \cdot \left( {\frac{Ri}{{Rh} + {Ri}} - \frac{Rk}{{Rj} + {Rk}}} \right)}} & {{Equation}\mspace{14mu} 11}\end{matrix}$

Setting the gain of the programmable gain instrumentation amplifier 302to a value G causes the output voltage V_(out) to be proportional to thevoltage difference between the measurement node 306 and the referencenode 309 (V_(i)−V_(k)), as illustrated in Equation 12:

V _(out) =G·(V _(i) −V _(k))   Equation 12

Substituting the expression in Equation 11 for (V_(i)−V_(k)) intoEquation 12 produces a relation between V_(out) and (V_(out)−V_(G)), asshown below in Equations 13 and 14:

$\begin{matrix}{V_{out} = {G \cdot \left( {V_{cc} - V_{G}} \right) \cdot \left( {\frac{Ri}{{Rh} + {Ri}} - \frac{Rk}{{Rj} + {Rk}}} \right)}} & {{Equation}\mspace{14mu} 13} \\{{\rho \equiv \frac{V_{out}}{G \cdot \left( {V_{cc} - V_{G}} \right)}} = \left( {\frac{Ri}{{Rh} + {Ri}} - \frac{Rk}{{Rj} + {Rk}}} \right)} & {{Equation}\mspace{14mu} 14}\end{matrix}$

When the ambient temperature is held constant and the effects ofself-heating are minimized (such that Rh does not change), resistancevalues in the bridge circuit will not change and the ratio p of theoutput voltage V_(out) to the gain G multiplied by the source-groundvoltage difference (V_(cc)−V_(G)) will be relatively constant. As shownin Equation 15, when ρ remains constant ΔV_(out)/(G·ΔV_(cc)) is alsoconstant:

$\begin{matrix}{\frac{\partial V_{out}}{\partial V_{cc}} = {G \cdot \rho}} & {{Equation}\mspace{14mu} 15}\end{matrix}$

FIG. 10 a illustrates a flow chart describing a self-heating calibrationprocess 1000 a in accordance with another aspect of the presentinvention for identifying the voltage settings at whichΔV_(out)/(G·ΔV_(cc)) is constant, and wherein this information isutilized to calibrate the temperature measurement circuit againstself-heating. In some embodiments, the self-heating calibration process1000 a may be performed by a self-heating calibration controller. In oneembodiment, this process involves setting the supply voltage V_(cc) at adesired voltage level V_(op) causing an unknown amount of self-heatingand an unknown temperature change ΔT_(op). The supply voltage V_(cc) isthen gradually lowered, which reduces the effect of self-heating.Because the temperature change ΔT is proportional to the square of thecurrent, the effects of self-heating will rapidly diminish as thevoltage V_(cc) is reduced. At a sufficiently low supply voltage V_(L),the effects of self-heating will be negligible. As described above, thelow supply voltage V_(L) at which self-heating effects are no longerobserved can be identified as the point at which the ratio ρ of theoutput voltage V_(out) to the gain G multiplied by the supply-sourcevoltage difference (V_(cc)−V_(G)) remains relatively constant despitechanges to the supply voltage V_(cc). At this point, a comparisonbetween the ratio ρ_(op) at the desired voltage level V_(op) with theratio ρ_(L) at the low voltage level V_(L) may be used to determine theeffect ΔV_(sh) of self-heating at the desired source voltage level usingthe relation shown in Equation 16:

ΔV _(sh) =G·(ρ_(op) −ρ _(L))·(V _(op) 31 V _(G))   Equation 16

As shown in FIG. 10 a, the self-heating calibration process 1000 a maybegin at step 1002, where the self-heating calibration controller setsthe source voltage V_(cc) to be equal to a desired operational voltageV_(op) such as, for example, the voltage that will be desired to providea good common mode signal and the ability to rapidly heat the RTD.

At step 1003, the self-heating calibration controller sets the gain Gfor the programmable gain instrumentation amplifier 302 to a desiredoperational gain value G_(op) such as, for example, the gain value thatwill maximize the signal-to-noise ratio of the temperature measurementcircuit 300.

After these initial values are set, at step 1004, the self-heatingcalibration controller measures the output voltage V_(out) of thetemperature measurement circuit 300.

At step 1005, the self-heating calibration controller calculates a ratioRatio_(op) between the measured output voltage V_(out) and theoperational supply-ground voltage (V_(cc)−V_(G)) multiplied by theoperational gain G_(op).

After the initial ratio Ratio_(op) is calculated, at step 1006, theself-heating calibration controller decreases the source voltage V_(cc).After the source voltage V_(cc) has been decreased, the self-heatingcalibration controller measures the new output voltage V_(out) at step1007 and calculates a new ratio Ratio_(i) at step 1008.

At step 1009, the self-heating calibration controller determines thedifference between the newly calculated ratio Ratio_(i) and thepreviously calculated ratio Ratio_(i-1). In the case that the differencebetween these ratios is above a predetermined threshold, that is,decreasing the source voltage V_(cc) continues to have a substantialeffect on the ratio, the self-heating calibration controller will returnto step 1006 and lower the source voltage V_(cc) again. In the case thatthe difference between these ratios is below the threshold (that is,decreasing the source voltage V_(cc) no longer has a substantial effecton the ratio), then the self-heating calibration controller hasidentified V_(L) and will proceed to step 1011.

At step 1011, the self-heating calibration controller calculates theself-heating voltage difference ΔV_(sh) by subtracting the ratioRatio_(i) calculated at the low voltage V_(L) from the ratio Ratio_(op)calculated at the operational voltage V_(op) and multiplying thedifference by the gain of the programmable gain instrumentationamplifier 302 and the desired operational source-ground voltage, inaccordance with Equation 16.

Referring to FIG. 11, a graph of the ratio ρ vs. the source voltageV_(cc) for an exemplary embodiment of the invention is illustrated. Ascan be seen in FIG. 11, at relatively high source voltages (e.g.V_(cc)=30 v), the slope of the graph is relatively higher, that is, ρ isnot constant. At relatively low source voltages (e.g. V_(cc)=10 v), theslope of the graph is substantially lower. At source voltages below 10volts, the value of the ratio p approaches a horizontal asymptote, thatis, ρ remains relatively constant. To calculate ΔV_(sh), the value of ρat the asymptote is subtracted from the value of ρ at the desiredoperational configuration.

FIG. 10 b illustrates a flow chart describing a self-heating calibrationprocess 1000 b in accordance with another embodiment of the presentinvention. As illustrated in FIG. 10 b, the self-heating process 1000 bis similar to self-heating process 1000 a. In this embodiment, at steps1004 b and 1007 b, the self-heating calibration controller measures thesource voltage V_(cc) using the same voltage measurement system as thatused to measure V_(out). This provides a real time measurement of thesupply voltage V_(cc) that can be used for the normalizationself-heating calibration process and may provide a more accurate valuefor V_(cc) than simply relying on an input voltage setting.

Referring to FIG. 10 c, a flow chart describing a self-heatingcalibration process 1000 c in accordance with another embodiment isillustrated. As illustrated in FIG. 10 c, the self-heating process 1000c is similar to self-heating process 1000 a. In this embodiment, at step1006 b-2, the self-heating calibration controller increases the gain Gof the programmable gain instrumentation amplifier 302. This increase,combined with the decrease of the source-ground voltage (V_(cc)−V_(G)),may be used to ensure a sufficient range of output voltages V_(out) evenwhen the source-ground voltage (V_(cc)−V_(G)) is small. In someembodiments, the gain G and the source voltage V_(cc) may be variedaccording to an inverse relationship such that the product of thesource-ground voltage (V_(cc)−V_(G)) and the gain G remains constant.For example, suppose the gain G is initially 10 and the source-groundvoltage (V_(cc)−V_(G)) is 30 v. A decrease in the source voltage V_(cc)of 5 volts (so the source-ground voltage is now 25 volts) should beaccompanied by an increase in the gain G from 10 to 12.

Thermal Calibration

With the chip 102, 202 loaded and the gross adjustments made to accountfor changes in heater resistance and changes in factors associated withchip loading, a fine thermal calibration may be desirable. Because PCRefficacy and diagnosis based on thermal melt depend heavily on theaccuracy of the temperature measurement, thermal calibration may berequired immediately before a diagnostic cycle begins. By measuring thevoltage response versus temperature or the implied resistance versustemperature relationship (where resistance may, for example, bedetermined based on Equation 4 above) the system can define a precisecalibration for the RTD immediately before a microfluidic chip 102, 202is used.

FIG. 12 illustrates a flow chart describing a thermal calibrationprocess 1200 in accordance with another aspect of this invention. Insome embodiments, the thermal calibration process 1200 may be performedby a thermal calibration controller. As shown in FIG. 12, process 1200may begin at step 1202, where the thermal calibration controller may setthe source voltage V_(cc) to the desired operational voltage V_(op).

At step 1203, the thermal calibration controller may set the gain valueG for the programmable gain instrumentation amplifier 302 to a desiredoperational gain voltage G_(op). The values of V_(op) and G_(op) may beselected, for example, to match the values that will be used when themicrofluidic device 202 is performing PCR or high-resolution melt.

After the initial values have been set, the thermal calibrationcontroller brings the temperature detecting circuit 300 to apredetermined temperature T_(n) at step 1204. In some embodiments, thisis achieved by utilizing an externally controlled heating device (e.g. aPeltier device, a resistive heater, etc.).

At step 1205, the thermal calibration controller measures the outputvoltage V_(n). At step 1206, the thermal calibration controller storesthe values of the temperature T_(n) and the output voltage V_(n) arestored in association with each other.

Then, the thermal calibration controller 1200 returns to step 1204wherein the temperature detecting circuit 300 is brought to a newtemperature T_(n). This is repeated until the thermal calibrationcontroller measures and stores a predetermined number M of (T_(n),V_(n)) relationships.

With the stored set of (T_(n), V_(n)), the temperature control system120 can determine precise values for temperature detection. Theinterpolation may take the form of a curve with one or more constantsfor each resistive sensor on the platform chip (such as a 3 termquadratic calibration curves), or calibration may take the form of alook-up table with set voltages (or resistances) for each temperature.

In some embodiments, the externally controlled heating device is able togenerate a uniform temperature environment for the platform chip and isable to precisely measure temperature. The external temperaturemeasurement may be made by any suitable device including an RTD, athermocouple, a thermistor, a semiconductor junction device, etc. Theexternal temperature measurement device should be factory or third partycalibrated and its calibration data should be embedded in controlsoftware, which may be configured to include this calibration data aspart of thermal calibration process 1200.

Embodiments of the present invention have been fully described abovewith reference to the drawing figures. Although the invention has beendescribed based upon these preferred embodiments, it would be apparentto those of skill in the art that certain modifications, variations, andalternative constructions could be made to the described embodimentswithin the spirit and scope of the invention.

Additionally, while the process described above and illustrated in thedrawings is shown as a sequence of steps, this was done solely for thesake of illustration. Accordingly, it is contemplated that some stepsmay be added, some steps may be omitted, the order of the steps may bere-arranged, and some steps may be performed in parallel.

1. A tunable temperature measurement circuit comprising: a source nodemaintained at a predetermined source voltage; a ground node maintainedat a predetermined ground voltage; a bridge circuit comprising: a firstresistance temperature detector connected between the source node and afirst measurement node, a first reference resistor connected between thefirst measurement node and the ground node, a potentiometer connectedbetween the source node and a reference node, and a scaling resistorconnected between the reference node and the ground node; and a firstprogrammable gain instrumentation amplifier wherein a first input to thefirst programmable gain instrumentation amplifier is connected to thereference node, a second input to the first programmable gaininstrumentation amplifier is connected to the first measurement node,and the output of the first programmable gain instrumentation amplifieris representative of the temperature sensed by the first resistancetemperature detector.
 2. The tunable temperature measurement circuit ofclaim 1, wherein the potentiometer is a programmable digitalpotentiometer.
 3. The tunable temperature measurement circuit of claim1, further comprising a capacitor connected in parallel with the scalingresistor.
 4. The tunable temperature measurement circuit of claim 1,further comprising a low-pass filter coupled to the output of the firstprogrammable gain instrumentation amplifier.
 5. The tunable temperaturemeasurement circuit of claim 1, further comprising a bypass circuitconnected between the first measurement node and the ground node,wherein the bypass circuit comprises a bypass switch in series with abypass resistor.
 6. The tunable temperature measurement circuit of claim5, wherein the bypass switch comprises a digital switch.
 7. The tunabletemperature measurement circuit of claim 5, wherein the bypass circuitis configured to pulse width modulate a current passing through thefirst resistance temperature detector.
 8. The tunable temperaturemeasurement circuit of claim 1, further comprising a power controlcircuit connected to the first measurement node, wherein the powercontrol circuit comprises: a bottom power switch connected between themeasurement node and a bottom power node maintained at the predeterminedsource voltage; and a grounding switch connected in series with a bypassresistor between the measurement node and the ground node.
 9. Thetunable temperature measurement circuit of claim 8, further comprising ashunt circuit connected between the reference resistor and the groundnode, wherein the shunt circuit comprises a shunt switch in parallelwith a shunt resistor.
 10. The tunable temperature measurement circuitof claim 1, further comprising: a selector switch disposed in betweenthe first resistance temperature detector and the first measurementnode; and one or more second resistance temperature detectors connectedto the source node in parallel with the first resistance temperaturedetector; wherein the selector switch is configured to connect one ofthe first resistance temperature detector and the one or more secondresistance temperature detectors to the measurement node.
 11. Thetunable temperature measurement circuit of claim 1, further comprising:a second resistance temperature detector connected between the sourcenode and a second measurement node, a second reference resistorconnected between the second measurement node and the ground; and asecond programmable gain instrumentation amplifier wherein a first inputto the second programmable gain instrumentation amplifier is connectedto the reference node, a second input to the second programmable gaininstrumentation amplifier is connected to the second measurement node,and the output of the second programmable gain instrumentation amplifieris representative of the temperature sensed by the second resistancetemperature detector.
 12. The tunable temperature measurement circuit ofclaim 11, further comprising a unity gain buffer, wherein the referencenode is connected to the programmable gain instrumentation amplifiersvia the unity gain buffer.
 13. The tunable temperature measurementcircuit of claim 1, wherein one or more of the first reference resistorand the scaling resistor are also potentiometers.
 14. In a tunabletemperature measurement system comprising: a source node maintained at apredetermined source voltage; a ground node maintained at apredetermined ground voltage; a bridge circuit comprising: a firstresistance temperature detector connected between the source node and afirst measurement node, a first reference resistor connected between thefirst measurement node and the ground node, a potentiometer connectedbetween the source node and a reference node, and a scaling resistorconnected between the reference node and the ground node; and a firstprogrammable gain instrumentation amplifier wherein a first input to thefirst programmable gain instrumentation amplifier is connected to thereference node, a second input to the first programmable gaininstrumentation amplifier is connected to the first measurement node,and the output of the first programmable gain instrumentation amplifieris representative of the temperature sensed by the first resistancetemperature detector, a method of calibrating the potentiometercomprising the steps of: (a) setting the resistance value of thepotentiometer to a first resistance value; (b) setting the gain of thefirst programmable gain instrumentation amplifier to a first gain value;(c) measuring the voltage output from the first programmable gaininstrumentation amplifier; (d) in the case that the measured voltage isabove a predetermined target value, adjusting the resistance value ofthe potentiometer in a first direction; (e) in the case that themeasured voltage is below the predetermined target value, adjusting theresistance value of the potentiometer in a direction opposite to thefirst direction; and (f) repeating steps (c) through (e) until themeasured voltage from the first programmable gain instrumentationamplifier is equal to the predetermined target value.
 15. The method ofcalibrating the potentiometer in a tunable temperature measurementsystem of claim 14, wherein the predetermined target value is selectedto maximize the signal to noise ratio in the output of the firstprogrammable gain instrumentation amplifier.
 16. The method ofcalibrating the potentiometer in a tunable temperature measurementsystem of claim 12, further comprising the steps of: (g) afterperforming step (f), storing the resistance value of the potentiometerin an electronic memory; (h) associating the stored resistance valuewith an identifier corresponding to the first resistance temperaturedetector; (i) repeating steps (a) through (h) for a plurality ofresistance temperature detectors to create a plurality of associationsbetween resistance temperature detectors and resistance values; (j)detecting the presence of one of the plurality of resistance temperaturedetectors; and (k) setting the resistance value of the potentiometer tothe resistance value associated with the one of the plurality ofresistance temperature detectors.
 17. The method of calibrating thepotentiometer in a tunable temperature measurement system of claim 16,wherein the step of detecting the presence of one of the plurality ofresistance temperature detectors comprises reading a machine readablebar code from a platform chip containing the one of the plurality ofresistance temperature detectors.
 18. The method of calibrating thepotentiometer in a tunable temperature measurement system of claim 16,wherein the step of detecting the presence of one of the plurality ofresistance temperature detectors comprises reading an RFID tag from aplatform chip containing the one of the plurality of resistancetemperature detectors.
 19. In a tunable temperature measurement systemcomprising: a source node maintained at a predetermined source voltage;a ground node maintained at a predetermined ground voltage; a bridgecircuit comprising: a first resistance temperature detector connectedbetween the source node and a first measurement node, a first referenceresistor connected between the first measurement node and the groundnode, a potentiometer connected between the source node and a referencenode, and a scaling resistor connected between the reference node andthe ground node; and a first programmable gain instrumentation amplifierwherein a first input to the first programmable gain instrumentationamplifier is connected to the reference node, a second input to thefirst programmable gain instrumentation amplifier is connected to thefirst measurement node, and the output of the first programmable gaininstrumentation amplifier is representative of the temperature sensed bythe first resistance temperature detector, a method of calibrating theself-heating properties of the tunable temperature measurement systemcomprising the steps of: (a) setting the predetermined source voltage toa first source voltage value corresponding to a desired operationalsupply voltage; (b) setting the gain of the first programmable gaininstrumentation amplifier to a first gain value corresponding to adesired operational gain value; (c) measuring the voltage output fromthe first programmable gain instrumentation amplifier; (d) determining afirst ratio of the output from the first programmable gaininstrumentation amplifier to the source node voltage multiplied by thegain of the first programmable gain instrumentation amplifier; (e)decreasing the predetermined source voltage to a new source voltagevalue; (f) measuring the voltage output from the first programmable gaininstrumentation amplifier; (g) determining a new ratio of the outputfrom the first programmable gain instrumentation amplifier to themeasured source node voltage multiplied by the gain of the firstprogrammable gain instrumentation amplifier; (h) determining anasymptote ratio by repeating steps (e) through (g) until the change ofthe new ratio determined at (g) between subsequent iterations is beneatha predetermined threshold; and (i) determining an operationalself-heating voltage difference by multiplying the desired operationalgain value by the source voltage and the difference between the firstratio and the asymptote ratio.
 20. The method of calibrating theself-heating properties of a tunable temperature measurement system ofclaim 19, wherein: step (c) further comprises measuring the voltage atthe source node; step (f) further comprises measuring the voltage at thesource node; and steps (d), (g), and (i) use the measured voltage at thesource node as the source node voltage.
 21. The method of calibratingthe self-heating properties of a tunable temperature measurement systemof claim 19, wherein step (e) further comprises increasing the gain ofthe first programmable gain instrumentation amplifier to a new gainvalue such that the product of the first source voltage value and thefirst gain value is equal to the product of the new source voltage valueand the new gain value.
 22. In a tunable temperature measurement systemcomprising: a source node maintained at a predetermined source voltage;a ground node maintained at a predetermined ground voltage; a bridgecircuit comprising: a first resistance temperature detector connectedbetween the source node and a first measurement node, a first referenceresistor connected between the first measurement node and the groundnode, a potentiometer connected between the source node and a referencenode, and a scaling resistor connected between the reference node andthe ground node; and a first programmable gain instrumentation amplifierwherein a first input to the first programmable gain instrumentationamplifier is connected to the reference node, a second input to thefirst programmable gain instrumentation amplifier is connected to thefirst measurement node, and the output of the first programmable gaininstrumentation amplifier is representative of the temperature sensed bythe first resistance temperature detector, a method for performingthermal calibration of the tunable temperature measurement systemcomprising the steps of: (a) setting the predetermined source voltage toa desired operational supply voltage; (b) setting the gain of the firstprogrammable gain instrumentation amplifier to a desired operationalgain value; (c) bringing the resistance temperature detector to a knowntemperature; (d) measuring a voltage output from the first programmablegain instrumentation amplifier; (e) storing the measured output voltagein an electronic memory in association with the known temperature; (f)repeating steps (c) through (e) to store a plurality of associationsbetween known temperatures and corresponding measured output voltages;and (g) utilizing the stored associations to calibrate the circuit forthermal variations.
 23. The method for performing thermal calibration ofa tunable temperature measurement system of claim 22, wherein the stepof bringing the resistance temperature detector to a known temperaturecomprises utilizing an externally controlled heating device that hasbeen independently calibrated.
 24. The method for performing thermalcalibration of a tunable temperature measurement system of claim 23,wherein the externally controlled heating device comprises a Peltierdevice.
 25. The method for performing thermal calibration of a tunabletemperature measurement system of claim 23, wherein the externallycontrolled heating device comprises a resistive heater.
 26. The methodfor performing thermal calibration of a tunable temperature measurementsystem of claim 22, wherein the step of utilizing the storedcorrelations comprises utilizing a look up table for the plurality ofknown temperatures.
 27. The method for performing thermal calibration ofa tunable temperature measurement system of claim 22, wherein the stepof utilizing the stored correlations comprises calculating a suitablecurve to interpolate output voltage between the known temperatures. 28.A system of controlling the temperature of a microfluidic device forperforming biological reactions, comprising tunable temperaturemeasurement circuit comprising: a source node maintained at apredetermined source voltage; a ground node maintained at apredetermined ground voltage; a bridge circuit comprising: a firstresistance temperature detector connected between the source node and afirst measurement node, a first reference resistor connected between thefirst measurement node and the ground node, a potentiometer connectedbetween the source node and a reference node, and a scaling resistorconnected between the reference node and the ground node; and a firstprogrammable gain instrumentation amplifier wherein a first input to thefirst programmable gain instrumentation amplifier is connected to thereference node, a second input to the first programmable gaininstrumentation amplifier is connected to the first measurement node,and the output of the first programmable gain instrumentation amplifieris representative of the temperature sensed by the first resistancetemperature detector.
 29. The system of controlling the temperature of amicrofluidic device for performing biological reactions of claim 28,wherein the potentiometer is a programmable digital potentiometer. 30.The system of controlling the temperature of a microfluidic device forperforming biological reactions of claim 28, further comprising acapacitor connected in parallel with the scaling resistor.
 31. Thesystem of controlling the temperature of a microfluidic device forperforming biological reactions of claim 28, further comprising alow-pass filter coupled to the output of the first programmable gaininstrumentation amplifier.
 32. The system of controlling the temperatureof a microfluidic device for performing biological reactions of claim28, further comprising a bypass circuit connected between the firstmeasurement node and the ground node, wherein the bypass circuitcomprises a bypass switch in series with a bypass resistor.
 33. Thesystem of controlling the temperature of a microfluidic device forperforming biological reactions of claim 32, wherein the bypass switchcomprises a digital switch.
 34. The system of controlling thetemperature of a microfluidic device for performing biological reactionsof claim 32, wherein the bypass circuit is configured to pulse widthmodulate a current passing through first resistance temperaturedetector.
 35. The system of controlling the temperature of amicrofluidic device for performing biological reactions of claim 28,further comprising a power control circuit connected to the firstmeasurement node, wherein the power control circuit comprises: a bottompower switch connected between the measurement node and a bottom powernode maintained at the predetermined source voltage; and a groundingswitch connected in series with a bypass resistor between themeasurement node and the ground node.
 36. The system of controlling thetemperature of a microfluidic device for performing biological reactionsof claim 35, further comprising a shunt circuit connected between thereference resistor and the ground node, wherein the shunt circuitcomprises a shunt switch in parallel with a shunt resistor.
 37. Thesystem of controlling the temperature of a microfluidic device forperforming biological reactions of claim 28, further comprising: asecond resistance temperature detector connected between the source nodeand a second measurement node, a second reference resistor connectedbetween the second measurement node and the ground; and a secondprogrammable gain instrumentation amplifier wherein a first input to thesecond programmable gain instrumentation amplifier is connected to thereference node, a second input to the second programmable gaininstrumentation amplifier is connected to the second measurement node,and the output of the second programmable gain instrumentation amplifieris representative of the temperature sensed by the second resistancetemperature detector.
 38. The system of controlling the temperature of amicrofluidic device for performing biological reactions of claim 37,further comprising a unity gain buffer, wherein the reference node isconnected to the programmable gain instrumentation amplifiers via theunity gain buffer.
 39. The system of controlling the temperature of amicrofluidic device for performing biological reactions of claim 28,wherein one or more of the first reference resistor and the scalingresistor are also potentiometers.
 40. The system of controlling thetemperature of a microfluidic device for performing biological reactionsof claim 28, further comprising a bridge adjustment controllerconfigured to: (a) set the resistance value of the potentiometer to afirst resistance value; (b) set the gain of the first programmable gaininstrumentation amplifier to a first gain value; (c) measure the voltageoutput from the first programmable gain instrumentation amplifier; (d)in the case that the measured voltage is above a predetermined targetvalue, adjust the resistance value of the potentiometer in a firstdirection; (e) in the case that the measured voltage is below thepredetermined target value, adjust the resistance value of thepotentiometer in a direction opposite to the first direction; and (f)repeat steps (c) through (e) until the measured voltage from the firstprogrammable gain instrumentation amplifier is equal to thepredetermined target value.
 41. The system of controlling thetemperature of a microfluidic device for performing biological reactionsof claim 40, wherein the bridge adjustment controller is furtherconfigured to: (g) store the resistance value of the potentiometer in anelectronic memory after performing step (f); (h) associate the storedresistance value with an identifier corresponding to the firstresistance temperature detector; (i) repeat steps (a) through (h) for aplurality of resistance temperature detectors to create a plurality ofassociations between resistance temperature detectors and resistancevalues;
 42. The system of controlling the temperature of a microfluidicdevice for performing biological reactions of claim 40, furthercomprising a self-heating calibration controller configured to: (a) setthe predetermined source voltage to a first source voltage valuecorresponding to a desired operational supply voltage; (b) set the gainof the first programmable gain instrumentation amplifier to a first gainvalue corresponding to a desired operational gain value; (c) measure thevoltage output from the first programmable gain instrumentationamplifier; (d) determine a first ratio of the output from the firstprogrammable gain instrumentation amplifier to the source node voltagemultiplied by the gain of the first programmable gain instrumentationamplifier; (e) decrease the predetermined source voltage to a new sourcevoltage value; (f) measure the voltage output from the firstprogrammable gain instrumentation amplifier; (g) determine a new ratioof the output from the first programmable gain instrumentation amplifierto the measured source node voltage multiplied by the gain of the firstprogrammable gain instrumentation amplifier; (h) determine an asymptoteratio by repeating steps (e) through (g) until the change of the newratio determined at (g) between subsequent iterations is beneath apredetermined threshold; and (i) determine an operational self-heatingvoltage difference by multiplying the desired operational gain value bythe source voltage and the difference between the first ratio and theasymptote ratio.
 43. The system of controlling the temperature of amicrofluidic device for performing biological reactions of claim 42,wherein the self-heating calibration controller is further configuredto: increase the gain of the first programmable gain instrumentationamplifier to a new gain value during step (e) such that the product ofthe first source voltage value and the first gain value is equal to theproduct of the new source voltage value and the new gain value.
 44. Thesystem of controlling the temperature of a microfluidic device forperforming biological reactions of claim 42, further comprising athermal calibration controller configured to: (a) set the predeterminedsource voltage to a desired operational supply voltage; (b) set the gainof the first programmable gain instrumentation amplifier to a desiredoperational gain value; (c) bring the resistance temperature detector toa known temperature; (d) measure a voltage output from the firstprogrammable gain instrumentation amplifier; (e) store the measuredoutput voltage in an electronic memory in association with the knowntemperature; (f) repeat steps (c) through (e) to store a plurality ofassociations between known temperatures and corresponding measuredoutput voltages; and (g) utilize the stored associations to calibratethe circuit for thermal variations.